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1. INTRODUCTION
There are two major causes of high switching stress in the flyback converter. Both are associated with the turn-off behavior of a bipolar transistor with an inductive load. The most obvious effect is the tendency for the collector voltage to over-shoot during turn-off, caused mainly by the transformer leakage inductance. The second, less obvious effect is the high secondary breakdown stress that will occur during turn-off if load line shaping is not used.
The voltage overshoot problem is best dealt with by ensuring that the leakage inductance is as small as possible, then clamping the tendency to overshoot by dissipative or energy recovery methods. The following section describes a dissipative clamp system. A more efficient energy recovery method using an extra winding is described in Part 2, Sec. 8.5.
If the energy recovery winding method is to be used with the flyback converter, the clamp voltage should be at least 30% higher than the reflected secondary voltage, to ensure efficient transfer of energy to the secondary. (The extra flyback voltage is required to drive current more rapidly through the secondary leakage inductance).
2. SELF-TRACKING VOLTAGE CLAMP
When a transistor in a circuit with an inductive or transformer load is turned off, the collector will tend to fly to a high voltage as a result of the energy stored in the magnetic field of the inductor or leakage inductance of the transformer.
In most flyback converters, the majority of the energy stored in the transformer will be transferred to the secondary during the flyback period. However, because of the leakage inductance, there will still be a tendency for the collector voltage to overshoot at the beginning of the flyback period unless some form of voltage clamp is provided.
In Fig. 1, the cumulative effects of transformer leakage inductance, the inductance of output capacitor, and loop inductance of the secondary circuit have been lumped together as LLT and referred to the primary side of the transformer in series with the main primary inductance Lp.
Fig. 1 (a) Stress-reducing self-tracking collector voltage clamp applied
to the primary of a flyback converter. (b) Collector voltage waveform, showing
voltage clamp action.
Consider the action during turn-off following an "on" period during which a current has been established in the primary winding of T1. When transistor Q1 turns off, all transformer winding voltages will reverse by flyback action. The secondary voltage Vs will not exceed the output Vc, except by the output rectifier D1 diode drop. The collector of Q1 is partly isolated from this clamp action by the leakage inductance LLT, and the energy stored in LLT will take the collector voltage more positive.
If the clamp circuit, D2, C2, were not provided, then this flyback voltage could be damagingly high, as the energy stored in LLC would be redistributed into the leakage capacitance seen at the collector of Q1.
However, in Fig. 1, under steady-state conditions, the required clamping action is provided by components D2, C2, and R1, as follows.
C2 will have been charged so that its plus end is at a voltage slightly more positive than the reflected secondary flyback voltage. When Q1 turns off, the collector voltage will fly back to this value, at which point diode D2 will conduct and hold the voltage constant (C2 being large compared with the captured energy). At the end of the clamping action, the voltage on C2 will be somewhat higher than its starting value.
During the remainder of the cycle, the voltage on C1 will return to its original value as a result of the discharge current flowing in R1. The spare flyback energy is thus dissipated in R1. This clamp voltage is self-tracking, as the voltage on C2 will automatically adjust its value, under steady-state conditions, until all the spare flyback energy is dissipated in R1.
If all other conditions remain constant, the clamp voltage may be reduced by reducing the value of R1 or the leakage inductance LLT.
It is undesirable to make the clamp voltage too low, as the flyback overshoot has a useful function. It provides additional forcing volts to drive current into the secondary leak age inductance during the flyback action. This results in a more rapid increase in flyback current in the transformer secondary, improving the transfer efficiency and reducing the losses incurred in R1. This is particularly important for low-voltage, high-current outputs, where the leakage inductance is relatively large. Therefore, it is a mistake to choose too low a value for R1, and hence a low clamp voltage. The maximum permitted primary voltage overshoot will be controlled by the transistor VCEX rating and should not be less than 30% above the reflected secondary voltage. If necessary, use fewer secondary turns.
If the energy stored in LLT is large and excessive dissipation in R1 is to be avoided, this network may be replaced by an energy recovery winding and diode, as would be used in a forward converter. This will return the spare flyback energy to the supply.
It should be clear that for high efficiency and minimum stress on Q1, the leakage inductance LLT should be made as small as possible. This will be achieved by good interleaving of the primary and secondary of the transformer. It is also necessary to choose minimum inductance in the output capacitor, and, most important, minimum loop inductance in the secondary circuits. The latter may be achieved by keeping wires from the transformer as closely coupled as possible and ideally twisted, running the tracks on the printed circuit board as a closely coupled parallel pair, and keeping distances small. It is attention to these details that will provide high efficiency, good regulation, and good cross regulation in the flyback-mode power supply.
3. FLYBACK CONVERTER "SNUBBER" NETWORKS
The turn-off secondary breakdown stress problem is usually dealt with by "snubber networks"; a typical circuit is shown in Fig. 2. The design of the snubber network is more fully covered in Part 1, Chap. 18.
Snubber networks will be required across the switching transistor in off-line flyback converters to reduce secondary breakdown stress. Also, it is often necessary to snub rectifier diodes to reduce the switching stress and RF radiation problems.
Fig. 2 Dissipative "snubber" circuit applied to the collector
of an off-line flyback converter.
In Fig. 2, snubber components Ds, Cs, and Rs are shown fitted across the collector and emitter of Q1 in a typical flyback converter. Their function is to provide an alternative path for the inductively driven primary current and reduce the rate of change of voltage (dv/dt) on the collector of Q1 during the turn-off action of Q1.
The action is as follows: As Q1 starts to turn off, the voltage on its collector will rise.
The primary current will now be diverted via diode Ds into capacitor Cs. Transistor Q1 turns off very quickly, and the dv/dt on the collector will be defined by the original collector cur rent at turn-off and the value of Cs.
The collector voltage will now ramp up until the clamp value (2Vcc) is reached. Shortly after this, because of leakage inductance, the voltage in the output secondary winding will have risen to Vsec (equal to the output voltage plus a diode drop), and the flyback current will be commutated from the primary to the secondary via D1 to build up at a rate con trolled by the secondary leakage inductance.
In practice, Q1 will not turn off immediately, and if secondary breakdown is to be avoided, the choice of snubber components must be such that the voltage on the collector of Q1 will not exceed Vceo before the collector current has dropped to zero, as shown in Fig. 3.
Fig. 3 Collector voltage and collector current waveforms, showing phase
shift when dissipative snubber components are fitted. Also snubber current
waveform during Q1 turn-off.
Unless the turn-off time of Q1 is known, the optimum choice for these components is an empirical one, based upon measurements of collector turn-off voltage and current. Part 1, Chap. 18 and Fig. 1.18.2a, b, c, and d show typical turnoff waveforms and switching stress with and without snubber networks.
A safe voltage margin should be provided on the collector voltage when the current is zero, say at least 30% below Vceo, as there is a considerable influence on these parameters from operating temperatures, loads, the spread of transistor parameters and the drive design. Figure 2.3.3 shows the limiting condition; in this example, the collector current has just dropped to zero as the collector voltage "hits" Vceo.
On the other hand, too large a value of Cs should be avoided, since the energy stored in this capacitor at the end of the flyback period must be dissipated in Rs during the first part of the "on" period.
The value of Rs is a compromise selection. A very low resistance results in excessive current in Q1 during turn-on and will result in excessive dissipation during the "on" transient. Too large a resistance, on the other hand, will not provide sufficient discharge of Cs during the minimum "on" period.
The values shown are a good compromise choice for the 100-W example. However, a careful examination of the voltage and current waveforms on the Q1 collector, under narrow pulse conditions, is recommended. The selection must always be a compromise for this type of snubber. The optimum selection of snubber components is more fully covered in Part 1, Chap. 18, and more effective snubber methods may be used that avoid a compromise. (See "The Weaving Low-Loss Snubber Diode," Part 1, Sec. 18.10.)
4. QUIZ
1. Why is the switching transistor particularly susceptible to high-voltage switching stress in the flyback converter?
2. Why does the flyback voltage often exceed the value that would be indicated by the turns ratio between the primary and secondary circuits?
3. Describe two methods used to reduce the high-voltage stress on the flyback switching element.
Also see: Our other Switching Power Supply Guide